The invention pertains to the frequency conversion of an RF signal transmitted for color-television receivers at the picture-and sound-carrier frequencies by the "third method" using a suitable frequency reversal which makes the frequencies of the picture carrier, the chrominance subcarrier, and at least one sound carrier appear transformed into the baseband. Such a frequency conversion is described in Offenlegungsschrift DE-A-33 13 867; the "third method" is described in a book by H. Meinke and F. W. Gundlach, "Taschenbuch der Hochfrequenztechnik", 2nd Edition, Berlin, 1962, pages 1497 to 1500.
To explain the problem to be solved by the invention, the prior art arrangement will first be described with the aid of FIGS. 1 to 4 in which.
FIG. 1 is a block diagram of an example of a prior art frequency conversion circuit;
FIG. 2 shows schematically the spectrum of the RF signal;
FIG. 3 shows schematically the spectrum of the composite signal transformed into the baseband, with double frequency utilization; and
FIG. 4 shows schematically the spectrum of the base signal in the usual baseband position.
The circuit of FIG. 1 corresponds to the circuit shown in Offenlegungsschrift DE-A-33 13 867 for effecting frequency conversion by central reversal.
In the prior art frequency conversion, the frequency of an oscillator ho, i. e. the mixing or conversion frequency fo, is the exact center frequency between the frequency of the received picture carrier bt and that of the received chrominance subcarrier ft'. Therefore, frequency conversion by such radio-frequency mixing is also called "central reversal". Through the radio-frequency mixing, the RF signal hf, starting from the frequency 0 Hz, is so transformed into the baseband that the frequency of the transformed picture carrier bt* and that of the transformed chrominance subcarrier ft* coincide. This frequency is half the frequency of the chrominance subcarrier ft of the composite color signal plus the sound signal, the base signal f. The RF signal hf is applied to the inputs of the first signal mixer s1 and the second signal mixer s2. Each of these signal mixer contains a two-input mixer circuit whose output is fed to a low-pass filter having a passband equal to half the RF signal bandwidth. The output of the low-pass filter is fed to an automatic gain control amplifier. The amplitude-controlled output signals are the output signals m1 and m2 of the first and second signal mixers s1 and s2, respectively.
The tunable oscillator ho generates the conversion signal fo, which is fed to the first signal mixer s1 direct, and to the second signal mixer s2 through the 90-degree phase shifter pad. The output m1 of the first mixer is fed to the first demodulator g1 and the first limiter b1. The output m2 of the second mixer is fed to the second demodulator g2 and the second limiter b2. The output of the first limiter b1 is the first limited mixer signals mb1, and that of the second limiter b2 the second limited mixer signals mb2. Each of the these two signals is applied to one of two inputs of the phase comparator pv. The output of the latter controls the function of the 90-degree phase shifter pd.
The first and second demodulators g1, g2 are synchronous demodulators to which the first demodulating signal d1 and the second demodulating signal d2 are applied as control signals. The demodulating signals come from the burst-signal-processing circuit b, with the second demodulating signal d2 shifted in phase with respect to the first demodulating signal d1 by 90.degree..
The output of the burst-signal-processing circuit b is fed as a reference signal vb, which has half the chrominance-subcarrier frequency referred to the usual baseband, to one input of the first frequency comparator fv. The other input of the latter is presented with the first limited mixer signal mb1, and the comparator output, the control signal fs, controls the frequency of the oscillator ho.
The outputs of the first demodulator g1 and the second demodulator g2 are applied to the adder ad, whose output is fed to the video low-pass filter vt. The output of the latter is the composite color signal plus the sound signal, the base signal f, in the usual baseband position. The passband of the video low-pass filter vt is equal to the bandwidth of the base signal f.
The base signal f is fed to the input of the burst-signal-processing circuit b, which generates the chrominance subcarrier ft. This circuit contains the chrominance-subcarrier filter, the variable-frequency chrominance-subcarrier oscillator, which oscillates at the frequency of the chrominance subcarrier, and a frequency divider which halves the frequency of the chrominance subcarrier ft; the output of the frequency divider is the reference signal vb and the first demodulating signal d1.
FIG. 2 shows the spectrum of the received RF signal hf, normalized to the frequency of the picture carrier bt. The actual video band with the picture carrier bt extends up to the upper video-band limit 31' at 5 MHz. Shown below the picture-carrier frequency is the limit of the transmitted vestigial sideband 30' at -1.25 MHz.
Outside the upper video-band limit 31, the first radio-frequency sound carrier tt1 is a 5.5 MHz, and the second radio-frequency sound carrier tt2' at 5.74 MHz. Within the video band, the color-signal transmission range extends from 5 MHz to 3.25 MHz. The radio-frequency chrominance subcarrier ft' is shown at 4.4. MHz for simplicity. Correspondingly, the signal fo of the oscillator ho is shown at 2.2 MHz.
FIG. 3 shows the composite signal obtained by the mixing processes in the first and second signal mixers s1, s2, which has been transformed into the baseband and shows a double utilization of the frequency range from 0 Hz. The fact that the bandwidth of this transformed composite signal is equal to only half the bandwidth of the RF signal hf can also be seen from the absolute scale, which is shown from 0 MHz to 3 MHz. The left-hand half of the spectrum of FIG. 2, i.e., the half below the frequency of the oscillator signal fo, thus appears reversed about this frequency toward positive values, so to speak. The second scale, given in round brackets, therefore relates to the left-hand portion of the spectrum, and the third scale, given in square brackets, to the original normalization of FIG. 2, i.e., the normalization to the picture carrier bt.
The frequency spectrum shows that the frequency of the transformed picture carrier bt* and that of the transformed chrominance subcarrier ft* coincide at 2.2 MHz. The transformed sound carrier tt1* still lies within the transmitted vestigial sideband, while the transformed sound carrier tt2* is just outside the vestigial sideband.
FIG. 4 shows schematically the spectrum of the base signal f, i.e., the composite color signal plus the sound signal, in the usual baseband. The picture carrier is identical with the frequency 0 Hz. The chrominance subcarrier ft is at 4.4 MHz--the above simplification applies here, too. The upper video-band limit 31 is at 5 MHz. Above this limit are the two sound carriers tt1, tt2.
The reversal of the double frequency utilization by the second frequency reversal at half the chrominance-subcarrier frequency result in the spurious signal ss at 4.4 MHz unless the second reversal is effected so that the addition by means of the adder ad cause total cancellation of the inverse spectrum; in practice, therefore, the spurious frequency ss must always be allows for. As a result of the second reversal, the vestigial sideband, which is still present in FIG. 3, has disappeared.
The aforementioned coincidence of the transformed picture carrier bt* and the transformed chrominance subcarrier ft* is to prevent any moire on the television screen as a result of the difference frequency between the transformed picture carrier bt* and the transformed chrominance subcarrier ft* if the two branches of the receiver with the two radio-frequency mixzers are not absolutely identical. To achieve this aim of absolute agreement between these specific frequencies, the oscillator ho must be locked exactly to the selected conversion frequency fo. However, in the case of television signals, which have carrier frequencies of up to 900 MHz, this is an almost unrealizable requirement because the phase-locked loop must respond so fast that any variations in the frequency of the oscillator ho are immediately compensated for. It is practically impossible to achieve the required synchronization if the oscillator ho produces frequency modulation noise. Even if its frequency deviation is very small, its phase deviation at 900 MHz is greater than 360 degrees. Theoretically, this frequency modulation noise can be counteracted only with extremely broadband control loops. These, however, are highly unstable and, thus, cannot be realized with sufficient reliability.